Power converter with winding communication

ABSTRACT

A power converter includes an energy transfer element, a switched element, a secondary control circuit, a primary switching, and a primary control circuit. The secondary control circuit generates a voltage pulse across a secondary winding of the energy transfer element while the secondary winding provides current to an output. The secondary control circuit is coupled to vary a voltage across the switched element to generate the voltage pulse across the secondary winding in response to an output of the power converter. The primary control circuit is coupled to the primary switch and a third winding of the energy transfer element. The primary control circuit is coupled to switch the primary switch to regulate the output in response to the voltage pulse. The secondary winding is coupled to reflect the voltage pulse onto the third winding which is on the same side of the energy transfer element as a primary winding.

TECHNICAL FIELD

This disclosure relates generally to power converters and, morespecifically, to ac-dc and/or dc-dc switched mode power converters.

BACKGROUND INFORMATION

Many electrical devices, such as cell phones, personal digitalassistants (PDA's), laptops, etc., use power to operate. Because poweris generally delivered through a wall socket as high voltage alternatingcurrent (ac), a device, typically referred to as a power converter, canbe used to transform the high voltage ac input to a well regulateddirect current (dc) output through an energy transfer element. Switchedmode power converters are commonly used due to their high efficiency,small size, and low weight to power many of today's electronics. In atypical operation, a switched mode power converter uses a switch toprovide the desired output quantity by either varying the duty ratio(typically the ratio of the on-time of the switch to the total switchingperiod) or by varying the switching frequency of the switch. Among thevarious switched mode power converter topologies, a flyback converter isa commonly used topology for low-cost power converters. In a typicalapplication, the ac-dc power converter receives an input from anordinary ac electrical outlet. The output of the power converter istypically a dc voltage, but may be a regulated dc current forapplications such as charging batteries.

Safety agencies generally require the power converter to providegalvanic isolation between input and output. Galvanic isolation preventsdc current between input and output of the power converter. In otherwords, a dc voltage applied between an input terminal and an outputterminal of the power converter will produce no substantial dc currentbetween the input terminal and the output terminal of the powerconverter. The requirement for galvanic isolation may be a complicationthat contributes to the cost of the power converter.

A power converter with galvanic isolation maintains an isolation barrierthat electrically separates the input from the output. Energy istransferred across the isolation barrier to provide power to the output,and information in the form of signals is transferred across theisolation barrier to regulate the output. Galvanic isolation istypically achieved with electromagnetic and electro-optical devices.Electromagnetic devices such as transformers and coupled inductors aregenerally used to transfer energy from input to output to provide outputpower, whereas electro-optical devices are generally used to transfersignals from output to input to control the transfer of energy frominput to output.

Efforts to reduce the cost of the power converter have focused on theelimination of electro-optical devices and their associated circuits.Alternative solutions generally use an energy transfer element such as atransformer or a coupled inductor to provide energy to the output andalso to obtain the information necessary to control the output. One lowcost configuration places the control circuit and a high voltage switchon the input side of the isolation barrier. The controller obtainsinformation about the output indirectly from observation of a voltage ata winding of the energy transfer element. The winding that provides theinformation is also on the input side of the isolation barrier.

The information about the output received in an indirect manner asdescribed above is based substantially on the magnetic coupling betweenthe windings placed on the input and the output sides of the energytransfer element. The magnetic coupling between the windings may not beperfect due to physical and mechanical limitations associated with theplacement of the windings of the energy transfer element. This may leadto inaccurate information about the output which may further lead topoor regulation of the output. Therefore, generating more reliableinformation about the output would improve regulation of power converteroutputs.

BRIEF DESCRIPTION OF THE DRAWINGS

Non-limiting and non-exhaustive embodiments of the invention aredescribed with reference to the following figures, wherein likereference numerals refer to like parts throughout the various viewsunless otherwise specified.

FIG. 1 is a functional block diagram of an example flyback powerconverter in accordance with the teachings of the present invention.

FIGS. 2A and 2B illustrate example representations of a switched elementin accordance with the teachings of the present invention.

FIG. 3 illustrates a functional block diagram of an example switchedelement in accordance with the teachings of the present invention.

FIG. 4 illustrates a functional block diagram of an example primarycontroller in accordance with the teachings of the present invention.

FIG. 5A illustrates a functional block diagram of an example secondarycontroller in accordance with the teachings of the present invention.

FIG. 5B illustrates a functional block diagram of another examplesecondary controller in accordance with the teachings of the presentinvention.

FIG. 6 illustrates a timing diagram of example waveforms of varioussignals of FIG. 1 when the power converter is operating in discontinuousconduction mode in accordance with the teachings of the presentinvention.

FIG. 7 illustrates a timing diagram of example waveforms of varioussignals of FIG. 1 when the power converter is operating in continuousconduction mode in accordance with the teachings of the presentinvention.

FIG. 8 is a flow diagram that illustrates an example process forregulating the output in accordance with the teachings of the presentinvention.

FIG. 9 is a flow diagram that illustrates an example process forregulating a power converter in accordance with the teachings of thepresent invention.

Corresponding reference characters indicate corresponding componentsthroughout the several views of the drawings. Skilled artisans willappreciate that elements in the figures are illustrated for simplicityand clarity and have not necessarily been drawn to scale. For example,the dimensions of some of the elements in the figures may be exaggeratedrelative to other elements to help to improve understanding of variousembodiments of the present invention. Also, common but well-understoodelements that are useful or necessary in a commercially feasibleembodiment are often not depicted in order to facilitate a lessobstructed view of these various embodiments of the present invention.

DETAILED DESCRIPTION

Embodiments of a power converter, a controller for a power converter,and a method of operating a power converter and a controller aredescribed herein. In the following description, numerous specificdetails are set forth to provide a thorough understanding of theembodiments. One skilled in the relevant art will recognize, however,that the techniques described herein can be practiced without one ormore of the specific details, or with other methods, components,materials, etc. In other instances, well-known structures, materials, oroperations are not shown or described in detail to avoid obscuringcertain aspects.

Reference throughout this specification to “one embodiment” or “anembodiment” means that a particular feature, structure, orcharacteristic described in connection with the embodiment is includedin at least one embodiment of the present invention. Thus, theappearances of the phrases “in one embodiment” or “in an embodiment” invarious places throughout this specification are not necessarily allreferring to the same embodiment. Furthermore, the particular features,structures, or characteristics may be combined in any suitable manner inone or more embodiments.

Particular features, structures or characteristics may be included in anintegrated circuit, an electronic circuit, a combinational logiccircuit, or other suitable components that provide the describedfunctionality. In addition, it is appreciated that the figures providedherewith are for explanation purposes to persons ordinarily skilled inthe art and that the drawings are not necessarily drawn to scale.

In a power converter with galvanic isolation, the input side of theisolation barrier of the power converter is sometimes referred to as theprimary side and the output side of the isolation barrier is sometimesreferred to as the secondary side. Windings of the energy transferelement that are not galvanically isolated from the primary side areprimary side windings, which are also sometimes called primaryreferenced windings. A winding on the primary side that is coupled to aninput voltage and receives energy from the input voltage is sometimesreferred to as the primary winding. Other primary referenced windingsthat deliver energy to circuits on the primary side may have names thatdescribe their principal function, such as for example a bias winding,or for example a sense winding. Windings that are galvanically isolatedfrom the primary side windings are secondary side windings, sometimescalled output windings.

A flyback power converter is a type of power converter that may providegalvanic isolation. Galvanic isolation is used for safety to preventharm to users from electrical contact with the ac power line. In orderto provide output regulation, the flyback power converter typically usesa switch on the primary side, often to referred to as a primary switch,and a primary side controller to control the primary switch to turn onor off in response to information about the output. The output may be avoltage, a current, or a combination of the two. Information about theoutput that the controller uses to regulate the output is generallyreferred to as feedback. Typically, a flyback converter may have acontrol algorithm to periodically enable (allow to turn on) and disable(prevent from turning on) the primary switch after a predefined time.The regular predefined period in which the primary switch may be turnedon or held off is typically known as a switching period or a switchingcycle. A switching cycle in which the primary switch is enabled may bereferred to as an enabled switching cycle. Typically, during an enabledswitching cycle the primary switch will be turned on and turned off. Theswitching of the primary switch in a predetermined switching cycle of aflyback converter may be disabled in response to a feedback signal. Aswitching cycle in which the primary switch is prevented from turning onmay sometimes be called as a skipped cycle. In general, there are twotypes of sensing schemes referred to as primary-side sensing andsecondary-side sensing which may provide the primary side controllerwith signals required to regulate the output of the power converter.With the secondary-side sensing, the primary controller may be coupledto a galvanically isolated device separate from the energy transferelement such as an optocoupler, to receive feedback information aboutthe output from a sensor coupled directly to the output on the secondaryside. In primary-side sensing, the sensor may include aprimary-referenced winding on the energy transfer element, typically abias winding that provides power to primary circuits, which is also usedto sense the secondary voltage indirectly to receive feedbackinformation about the output on the secondary side. A cost effectiveapproach to achieve superior regulation of the output at high efficiencymay use a primary-referenced switch controller with asecondary-referenced controller that senses the output directly.

The voltages on the windings of transformers and coupled inductors arerelated by the number of turns on each winding. The number of turns onthe primary winding may be referred to as N_(P) and the number of turnson the secondary winding may be referred to as N_(S). A voltage appliedat the primary winding generates a magnetic flux that couples with thesecondary winding to produce a voltage on the secondary winding. Thevoltage on the secondary winding is directly proportional to the numberof turns on the secondary winding and inversely proportional to thenumber of turns on the primary winding. Thus, the secondary windingvoltage is proportional to the primary winding voltage by the turnsratio of the windings.

The secondary winding is typically coupled to the load via a rectifier.The rectifier may be a passive electronic component such as a diode oran active electronic component such as a transistor. An active rectifierwhich uses a transistor is often referred to as a synchronous rectifier.Use of a synchronous rectifier may raise the efficiency of a powerconverter because a transistor may drop less voltage than a diode whenconducting the same current. Flyback power converters which usesynchronous rectifiers on the secondary side often include secondarycontrollers to turn on the transistor in the synchronous rectifier everytime energy is transferred to the secondary winding. In other words,every time there is current in the secondary winding, the secondarycontroller turns on the transistor in the synchronous rectifier.

As stated before, most primary side control schemes for power convertersobtain indirect information about the output voltage. These methods mayrely on a relationship between the voltage at a primary side winding andthe status of the output of the power converter. A difficulty with thismethod is that the relationship between the voltage at the primary sidewinding and the voltage at the output of the power converter is notprecisely known. The status of the output sensed by the primary sidewinding is typically an analog value resulting from the magneticcoupling between the primary side and secondary side windings. Althoughvoltages on primary side windings are approximately proportional to theoutput voltage of the power converter, many non-ideal effects such asimperfect magnetic coupling and voltage drops across conductingcomponents may lead to inaccurate information about the status of theoutput voltage. Any control scheme that is based on such type ofmeasurement and sensing of the status of the output may cause the outputto deviate from a desired regulated value. As stated before the outputmay be either a voltage, a current, or a combination of the two.Therefore, the term ‘desired regulated value’ may be interchangeablyused with ‘desired regulated voltage’ throughout the specification.

As will be discussed, a method and apparatus are disclosed with respectto an example flyback power converter, which provide a manner ofobtaining a two-state indication, or in other words, a ‘yes or no’indication of whether the output voltage is within a range of desiredregulated value. The disclosed method and apparatus avoid problems withimprecise indirect analog measurement. In one of the disclosedembodiments, every time the primary switch turns off and there is apositive current in the secondary winding, a momentary change in aprimary side winding voltage may be induced by operating a switchedelement on the secondary side in response to a secondary controller toalter the impedance of the switched element. The change in the impedanceof the switched element on the secondary side induces a momentary changein the secondary voltage while the secondary winding delivers current tothe output. This momentary change in the secondary voltage is reflectedas a momentary change in the bias winding voltage. The momentary changein the bias winding voltage provides a two-state indication, or in otherwords, a ‘yes or no’ indication to the primary controller whether theoutput is within a range of desired regulated value. In another example,the momentary change in the bias winding voltage may provide anindication about any other quantity which may represent the loadcondition, such as temperature or output current. In one example, theprimary controller may use this information about the output voltagealong with a control algorithm to decide whether to turn the primaryswitch on or to hold the primary switch off for the next switchingperiod or periods. In one example, the primary switch may be enabled(allowed to turn on) or disabled (prevented from turning on) for aportion of a predefined switching period or a switching cycle inaccordance with the teachings of the present invention. In yet anotherexample, the primary controller could incrementally adjust any otherparameter such as a duty ratio, on-time, off-time, switching frequency,or current limit.

In the example flyback power converter to be discussed, the switchedelement on the secondary side includes a synchronous rectifier. As willbe discussed, inducing a momentary change in secondary voltage isachieved by turning off a transistor in the synchronous rectifier on thesecondary side momentarily by a secondary controller when the primaryswitch turns off and the secondary winding is delivering current to theload.

More specifically, embodiments of the disclosure disclose a way toinduce an increase in the voltage at the secondary winding using thesecondary controller and observing the induced increase in the voltageat the bias winding in order to indicate to the primary controller ifthe output voltage is greater than or equal to the desired regulatedvalue.

In the example power converter, the secondary side controllermomentarily turns off the transistor in the synchronous rectifier on thesecondary side, to force a momentary increase in the bias windingvoltage when the output is greater than or equal to the desiredregulated value. The primary controller senses this momentary increasein the bias winding voltage as an indication that the output voltage isgreater than or equal to the desired regulated value.

In other words, by increasing the bias winding voltage in this way, theprimary controller may sense that the output voltage is greater than orequal to the desired regulated value. The disclosed method and apparatusmay be implemented as an additional circuitry to generate a precisecontrol signal for the primary switch to be turned on or held off.

As stated before, the illustrated embodiment to be discussed is aflyback power converter with a synchronous rectifier on the secondaryside. It is appreciated however, that the embodiments of the disclosuremay relate to any power converter topology where there is current in thesecondary winding when the primary switch is off.

FIG. 1 is a functional block diagram illustrating one example of a dc-dcpower converter 100 that receives an input voltage V_(IN) 102 to producean output voltage V_(O) 142 and an output current I_(O) 140 to a load128. In an example of an ac-dc power converter, the dc input voltageV_(IN) 102 may be a rectified and filtered ac input voltage. Inputvoltage V_(IN) 102 is positive with respect to an input return 120.Output voltage V_(O) 142 is positive with respect to an output return130. The example power converter 100 of FIG. 1 is a regulated flybackconverter. As shown, the power converter 100 includes energy transferelement T1 104, which in a flyback converter is a coupled inductor. Acoupled inductor is sometimes referred to as a transformer. From here onthroughout the specification the energy transfer element T1 104 may bereferred to as a transformer. Energy transfer element T1 104 isillustrated as having three windings, a primary winding 106 with N_(P)turns, a secondary winding 110 with N_(S) turns, and a bias winding 108with N_(B) turns. The voltages on the windings are related by the numberof turns on each winding. Secondary winding 110 of the transformer isgalvanically isolated from the primary winding 106 and bias winding 108.

The illustrated converter 100 further includes a primary switch S1 136,a primary control circuit 114, a clamp circuit 112, a first resistor R1116, a second resistor R2 118, a bias voltage signal 150, a switchedelement 122, a secondary control circuit 124, and an output capacitor C1126. Also shown in FIG. 1 are primary voltage V_(P) 156, primary currentI_(P) 132, a primary current sense signal 154, a primary drive signal152, a secondary current I_(S) 138, a secondary voltage V_(S) 148,secondary voltage sense signals 125 and 129, a secondary drive signal123, an input 162 of the switched element 122, an output 164 of theswitched element 122, a switched element voltage V_(SR) 160, a secondarycurrent sense signal 144, a bias winding voltage V_(B) 137, a biaswinding current I_(B) 158, a primary switch current I_(D) 134, an inputreturn 120, and an output return 130.

As shown in the depicted example, the primary switch S1 136 opens andcloses in response to the primary drive signal 152 from the primarycontrol circuit 114. In one example, primary switch S1 136 may be ametal oxide semiconductor field effect transistor (MOSFET). In anotherexample, primary switch S1 136 may be a bipolar junction transistor(BJT). In yet another example, primary switch S1 136 may be an insulatedgate bipolar transistor (IGBT) or other suitable switch. The primarycontrol circuit 114 and the primary switch S1 136 may be integrated.

In one example, primary control circuit 114 generates the primary drivesignal 152 in response to the bias voltage signal 150 to turn theprimary switch S1 136 on or to hold it off. The primary switch S1 136 isclosed when it is on. The primary switch S1 136 is open when it is off.Primary control circuit 114 may also be responsive to the primarycurrent sense signal 154 which indicates the value of primary switchcurrent I_(D) 134 in primary switch S1 136. Any of the several wayspracticed in the art to sense current in a switch may provide theprimary current sense signal 154. In one example, primary drive signal152 turns primary switch S1 136 off when primary current sense signal154 reaches a predetermined value.

The example power converter 100 of FIG. 1 also illustrates a clampcircuit 112 which is coupled across the primary winding 106. All of theenergy stored by the primary current I_(P) 132 through primary winding106 cannot be transferred to other windings because of imperfectmagnetic coupling between primary winding 106 and the other windings ofthe energy transfer element. In the example power converter 100 of FIG.1, energy that cannot be transferred to other windings is received bythe clamp circuit 112. The clamp circuit 112 is coupled across theprimary winding 106. The clamp circuit 112 limits the voltage acrossprimary winding 106 to protect the primary switch S1 136 from damage byexcessive voltage.

In one example, the primary control circuit 114 controls the switchingof the primary switch S1 136 with the primary drive signal 152 inresponse to a primary current sense signal 154 and a bias voltage signal150. The bias voltage signal 150 is the bias winding voltage V_(B) 137scaled by the resistors R1 116 and R2 118. In the depicted example,primary control circuit 114 controls the switching of the primary switchS1 136 to regulate the output of the power converter to the desiredregulated value. The output may be a voltage, a current, or acombination of a voltage and a current. The example power converter 100of FIG. 1 illustrates the primary control circuit 114 regulating theoutput voltage V_(O) 142. The load 128 receives the regulated outputvoltage V_(O) 142 and the output current I_(O) 140. When primary switchS1 136 is closed, the primary voltage V_(P) 156 is substantially equalto the input voltage V_(IN) 102 and there is primary current Ip 132 inthe primary winding 106 of the energy transfer element T1 104, storingenergy in the magnetic field of the energy transfer element T1 104.There is substantially no current in secondary winding 110 and in biaswinding 108 when primary switch S1 136 is closed. The switched element122 coupled to secondary winding 110 prevents current in the secondarywinding 110 when primary switch S1 136 is closed.

When the primary switch S1 136 is open, the primary voltage V_(P) 156 isthe negative of the reflected secondary voltage V_(S) 148 on thesecondary winding 110 owing to the magnetic coupling between the primarywinding 106 and the secondary winding 110. The secondary current I_(S)138 in the secondary winding 110 is non-zero once the primary switch S1136 opens. In the depicted example the resistors R1 116 and R2 118 haverelatively high impedance; therefore, current I_(B) 158 in the biaswinding is insubstantial when primary switch S1 136 opens. Thus, asubstantial portion of the energy stored in energy transfer element T1104 may be released through secondary winding 110 after primary switchS1 136 opens. That is, substantially all the energy stored by theprimary current I_(P) 132 in energy transfer element 104 when primaryswitch S1 136 is closed gets transferred to circuits which receive thesecondary current I_(S) 138 from the secondary winding 110 when theprimary switch S1 136 is open. The secondary current I_(S) 138 minus theoutput current I_(O) 140 charges the capacitor C1 126 to produce theoutput voltage V_(O) 142. In the example of FIG. 1, capacitor C1 126 hasa sufficient capacitance such that the output voltage V_(O) 142 issubstantially a dc voltage.

In the illustrated example, the approximate relationship between thebias winding voltage V_(B) 137 and voltage V_(S) 148 is determined bythe ratio of the number of turns of the respective windings 108 and 110.That is:

$\frac{V_{B}}{V_{S}} \simeq \frac{N_{B}}{N_{S}}$

As shown in the example illustrated in FIG. 1, power converter 100further includes a secondary control circuit 124 that operates aswitched element 122 to change the voltage V_(S) 148 on the secondarywinding 110 by altering the impedance of the switched element 122, whilesecondary winding 110 delivers the secondary current I_(S) 138 to theoutput in accordance with the teachings of the present invention. In theexample shown, the secondary control circuit 124 receives the voltageV_(S) 148 at secondary winding 110 as secondary voltage sense signals125 and 129. The secondary control circuit 124 receives the outputvoltage V_(O) 142 as signal 127. Secondary control circuit 124 producesa secondary drive signal 123 that controls the switched element 122 toalter the impedance of (and consequently the voltage across) theswitched element 122. By changing the impedance, the voltage V_(S) 148on the secondary winding 110 is altered momentarily in response to thedifference between the desired regulated voltage and the actual value ofoutput voltage V_(O) 142 when secondary winding 110 delivers thesecondary current I_(S) 138 to the output. The change in the biaswinding voltage V_(B) 137 across the bias winding 108 is observed byprimary control circuit 114 after scaling by the resistors R1 116 and R2118. The primary control circuit 114 controls the primary switch S1 136,in response to the change in the bias winding voltage V_(B) 137 tocontrol the primary switch S1 136 such that output voltage V_(O) 142 isregulated to the desired regulated voltage.

FIG. 2A illustrates a circuit model 200 of an example of the switchedelement 122 which includes a single pole double throw (SPDT) switchS_(A) 206. The SPDT switch S_(A) 206 may be controlled by the secondarydrive signal 123 to be in either position 1 or position 2. The secondarydrive signal 123 may have two different values, high or low, to set theposition of the switch S_(A) 206. When the secondary drive signal 123 ishigh, the switch S_(A) 206 is in position 1. When the secondary drivesignal 123 is low, the switch S_(A) 206 is in position 2.

FIG. 2A also shows an ideal diode 210. As such, the switched element 122is a unidirectional switch which may conduct current in only onedirection from the input 162 of the switched element 122 to the output164 of the switched element 122. In other words, the diode 210 preventsthe secondary current I_(S) 138 from becoming negative. In practice, thediode 210 is optional and not required, although it may be realized byany other electronic component that operates to make the switch S_(A)206 unidirectional.

When switch S_(A) 206 is in position 2, current passing between theinput 162 and the output 164 of the switched element 122 passes throughimpedance Z2 204. In addition, when switch S_(A) 206 is in position 2,the voltage between the input 162 and the output 164 of the switchedelement 122 is substantially equal to the voltage across the impedanceZ2 204. When switch S_(A) 206 is in position 1, current passing betweenthe input 162 of the output 164 of the switched element 122 passesthrough impedance Z1 202. Further, when switch S_(A) 206 is in position1, the voltage across the switched element 122 between the input 162 andthe output 164 of the switched element 122 is substantially equal to thevoltage across the impedance Z1 202.

In general, impedances Z1 202 and Z2 204 may be any suitable valueincluding zero, as long as they are different. Thus in the illustratedexample, high and low values of the secondary drive signal 123 offerdifferent impedance paths between input 162 and output 164 of theswitched element 122. In other words, the impedance of the switchedelement 122 is made higher or lower by switching between the impedancesZ1 202 and Z2 204. For a given current I_(S) 138 greater than zero, ahigher impedance of the switched element 122 causes an increase in theswitched element voltage V_(SR) 160, whereas a lower impedance of theswitched element 122 causes a decrease in the switched element voltageV_(SR) 160. Thus, a change in the impedance of the switched element 122may be used to change the switched element voltage V_(SR) 160. Further,the change in the switched element voltage V_(SR) 160 may be used tochange the secondary winding voltage V_(S) 148 in accordance with theteachings of the present invention.

FIG. 2B illustrates a circuit model 220 of another example switchedelement 122 that includes a single pole single throw (SPST) switch S_(B)226. The SPST switch S_(B) 226 is controlled by secondary drive signal123 to be either open or closed. When the secondary drive signal 123 ishigh, the switch S_(B) 226 is closed. When the secondary drive signal123 is low, the switch S_(B) 226 is open. FIG. 2B also includesimpedances Z3 222 and Z4 224. One difference between the exampleswitched element 122 shown in FIG. 2A and the example switched element122 shown in FIG. 2B is that in the example switched element 122 of FIG.2B, the impedance Z3 222 is between input 162 and output 164 of theswitched element 122 whether switch S_(B) 226 is open or closed. Whenswitch S_(B) 226 is open, substantially all current passing betweeninput 162 and output 164 of the switched element 122 passes throughimpedance Z3 222. There is substantially no current passing throughimpedance Z4 224. When switch S_(B) 226 is closed, the secondary currentI_(S) 138 passing between input 162 and output 164 of the switchedelement divides between the impedances Z3 222 and Z4 224.

FIG. 2B also shows an ideal diode 228. As such, the switched element 122is unidirectional and may conduct current only in one direction from theinput 162 to the output 164 of the switched element 122. The diode 228is optional and is not required. The function of the ideal diode may berealized by any other electronic component that operates to make theswitch S_(B) 226 unidirectional.

In general, impedances Z3 222 and Z4 224 may be any suitable value aslong as Z3 222 is not zero. Thus in the illustrated example, high andlow values of the secondary drive signal 123 offer higher and lowerimpedance paths between input 162 and output 164 of the switched element122. In other words, the impedance of the switched element 122 is madehigher or lower by switching between the impedances Z3 222 or Z3 222 inparallel with Z4 224. For a given current I_(S) 138 greater than zero, ahigher impedance of the switched element 122 causes an increase in theswitched element voltage V_(SR) 160, whereas, a lower impedance of theswitched element 122 causes a decrease in the switched element voltageV_(SR) 160. Thus, a change in the impedance of the switched element 122may be used to change the switched element voltage V_(SR) 160. Thechange in the switched element voltage V_(SR) 160 may be used to changethe secondary winding voltage V_(S) 148 in accordance with the teachingsof the present invention.

When switch S_(B) 226 is open, the secondary current I_(S) 138 passingbetween input 162 and output 164 of the switched element 122 of theswitched element 122 passes through impedance Z3 222. In addition, whenswitch S_(B) 226 is open, the voltage between input 162 and output 164of the switched element 122 is substantially equal to the voltage acrossthe impedance Z3 222. When switch S_(B) 226 is closed, the secondarycurrent I_(S) 138 passing between input 162 and output 164 of theswitched element 122 passes through the parallel combination ofimpedances Z3 222 and Z4 224, which is lower than the impedance Z3. Thuswith the switch open, the voltage between input 162 and output 164 ofthe switched element 122 is greater than the voltage when the switch isclosed for a given current I_(S) 138 greater than zero.

In one example, any of the impedances Z1 202, Z2 204, Z3 222, and Z4 224in FIG. 2A and FIG. 2B may be nonlinear. For example, a PN junctiondiode may be considered to have a nonlinear impedance.

FIG. 3 shows an example implementation 300 of the switched element 122in accordance with the teachings of the present invention. The switchedelement 122 comprises a transistor 306 and a diode 304 configured as asynchronous rectifier in FIG. 3. In one example the diode 304 mayrepresent the internal body diode of transistor 306. In another example,the diode 304 may be external to the transistor 306 and coupled in adirection which is parallel to the internal body diode of transistor306. The synchronous rectifier 122 is coupled such that diode 304 allowsthe secondary winding 110 to provide current I_(S) 138 to the output ofpower converter 100 of FIG. 1. As stated before, many topologies ofswitched mode power converters may use synchronous rectifiers on thesecondary side. The illustrated embodiment of the invention may beadapted to power converters that already use synchronous rectifierswithout much modification and thus provide a cost-effective way ofaccurate regulation.

The impedances Z1 202 shown in FIG. 2A and Z3 222 shown in FIG. 2B arerepresentatives of the forward characteristics of diode 304 in FIG. 3.The switch SB 226 in FIG. 2B is representative of the transistor 306 inFIG. 3. In one example, the impedance Z4 224 shown in FIG. 2B isrepresentative of the drain-source ON resistance (R_(DSON)) of thetransistor 306 in FIG. 3. The R_(DSON) of transistor 306 multiplied bythe current through the transistor 306 may be referred to as thetransistor voltage drop.

The secondary drive signal 123 controls the switching of the transistor306. When the secondary drive signal 123 is high with respect to theinput 162, the transistor 306 is on and when the secondary drive signal123 is low with respect to the input 162, the transistor 306 is off.

When the transistor 306 is on, the effective impedance between the input162 and the output 164 of the switched element 122 is equivalent to theR_(DSON) of the transistor 306 in parallel with the nonlinear forwardcharacteristic of the diode 304. However, as R_(DSON) is much less thanthe forward resistance of the diode 304, substantially all the currentpasses through the transistor 306. Therefore, the switched elementvoltage V_(SR) 160 when the transistor 306 is on is substantially equalto V_(T1), the voltage across the R_(DSON) of the transistor 306 whenthe secondary current I_(S) 138 is greater than zero.

When the transistor 306 is off, the effective impedance between theinput 162 and the output 164 of the switched element 122 is equivalentto the forward nonlinear characteristic of the diode 304. Therefore, theswitched element voltage V_(SR) 160 when the transistor 306 is off issubstantially equal to V_(D1), the forward voltage of the diode 304 whenthe secondary current I_(S) 138 is greater than zero.

As stated above, in the example power converter 100 of FIG. 1, theswitched element 122 switches the positive secondary current I_(S) 138between different impedance paths of Z1 202 and Z2 204 as shown in FIG.2A, or Z3 222 and Z4 224 in parallel with Z3 222 as shown in FIG. 2B. Inother words, the switched element 122 switches between a forwardnonlinear resistance of diode 304 or the R_(DSON) of the transistor 306in FIG. 3. The change in the impedance paths changes the switchedelement voltage V_(SR) 160 to either V_(T1) or V_(D1). The transistor306 is selected such that the voltage across the transistor 306 when thetransistor 306 conducts will be much less than the voltage across thediode 304 when the diode 304 conducts. Thus, when there is current inthe switched element 122, the switched element voltage V_(SR) 160 isequal to V_(T1) when transistor 306 is on and the switched elementvoltage V_(SR) 160 is equal to V_(D1) when the transistor 306 is off andthe diode 304 is conducting.

Referring to FIG. 1 and FIG. 3, the voltage across the secondary windingis given by:

V _(S) =V _(SR) +V _(O)

Therefore, the value of secondary voltage V_(S) 148 increases whentransistor 306 is off and the diode is conducting the positive secondarycurrent I_(S) 138. This increase in the value of secondary voltage V_(S)148 is reflected at the bias winding 108.

FIG. 4 illustrates a functional block diagram 400 of an example primarycontrol circuit 114 of the power converter 100 of FIG. 1 in accordancewith the teachings of the present invention. The primary control circuit114 includes a voltage observer circuit 402 and a drive circuit 404. Thevoltage observer circuit 402 is coupled to resistors R1 116 and R2 118that scale the bias winding voltage V_(B) 137 to a bias voltage signal150. The output 406 of the voltage observer circuit 402 is coupled to bereceived by the drive circuit 404. The drive circuit 404 outputs theprimary drive signal 152 which controls the turning on and off of theprimary switch S1 136. The voltage observer circuit 402 detects a changein the bias winding voltage V_(B) 137 whenever there is a change in thesecondary voltage V_(S) 148 while the secondary winding 110 isconducting positive current I_(S) 138. The change in the bias windingvoltage V_(B) 137 may result from the change in the switched elementvoltage V_(SR) 160. Referring to FIG. 1 and FIG. 4, this momentarychange in the bias winding voltage V_(B) 137 may provide an indicationthat the output voltage V_(O) 142 is greater than or equal to a desiredregulated voltage. In other words, the momentary change in the biaswinding voltage V_(B) 137 may indicate that the output is regulated. Thevoltage observer circuit 402 then provides an output 406 to the drivecircuit 404 which further either turns the primary switch S1 136 on andoff or holds it off in accordance with a control algorithm. Thus, theinformation about the status of the output voltage V_(O) 142 may be usedto operate the primary switch S1 136 as required to provide the desiredoutput from the power converter.

FIG. 5A is a functional block diagram 500A that illustrates details ofan example secondary control circuit 124 of the power converter 100 inaccordance with the teachings of the present invention. The secondarycontrol circuit 124 includes a synchronous rectifier controller 502, afirst two-input AND gate 520, a voltage comparator 504, a monostablemultivibrator circuit 510, an inverter 512, a second two-input AND gate514, and resistors R3 506 and R4 508. An output 522 of the synchronousrectifier controller 502 is coupled to a first input 522 of the AND gate520. AND gate 520 and monostable multivibrator 510 may be referred to asa pulse generator, while inverter 512 and AND gate 514 may be referredto as secondary logic.

Secondary control circuit 124 is coupled to vary switched elementvoltage V_(SR) 160 (which generates a voltage pulse across secondarywinding 110) in response to an output V_(O) 142 being at or above areference value (e.g. a target regulated dc voltage) while secondarycurrent I_(S) 138 is positive in the secondary winding 110. Thesynchronous rectifier controller 502 senses when the primary switch S1136 turns off through the secondary voltage V_(S) 148 and/or thesecondary current I_(S) 138 and outputs a delayed enable pulse at theoutput 522 after a predetermined delay. The delayed enable pulse at theoutput 522 of the synchronous rectifier controller 502, acts as atrigger for the monostable multivibrator circuit 510 when there is apositive current I_(S) 138 in the secondary winding 110. It isappreciated that the synchronous rectifier controller 502 may beconfigured to output a multiple number of pulses during a singleswitching cycle of the power converter 100 of FIG. 1. A second input 524of the AND gate 520 is coupled to the output of the voltage comparator504. The output 526 of the AND gate 520 is coupled to the input of themonostable multivibrator circuit 510. The output 532 of the monostablemultivibrator circuit 510 is coupled to the input of the inverter 512.Monostable multivibrator 510 generates a disable pulse at output 532when AND gate 520 receives delayed enable pulse 522 after a delay timeand output signal 524 is logic high. In one example, the delay time mayprovide information about output current, or temperature, or any otherquantity which may represent the load condition. In the example powerconverter 100 of FIG. 1, the delay time may be proportional to theoutput current. That is, the delay time may be longer for lower outputcurrent and shorter for higher output current. In other words, in theexample power converter 100 of FIG. 1, the delay time may be longer forlighter loads and shorter for heavier loads. In another embodiment thedelay time may be proportional to the temperature of the powerconverter. That is, the delay time may be longer for lower temperaturesand shorter for higher temperatures. In yet another embodiment the delaytime” may be proportional to another quantity indicating the loadcondition. An input of the AND gate 514 is coupled to receive an enablesignal 516 output by the synchronous rectifier controller 502. A secondinput of the AND gate 514 is coupled to the output 518 of the inverter512. The output of the AND gate 514 is coupled to generate secondarydrive signal 123. The inverting input of the voltage comparator 504 iscoupled to receive a reference voltage V_(REF) 528 which may be a scaledvalue of the desired regulated voltage V_(REG). The desired regulatedvoltage V_(REG) may be proportional to the reference voltage V_(REF) bya factor of K. In the depicted example, the relation between the outputvoltage V_(O) 142 and the desired regulated voltage V_(REG) 528 may begiven by:

V _(REF) =KV _(REG)

The value of K may be substantially given by:

$K = \frac{R\; 4}{{R\; 3} + {R\; 4}}$

A scaled quantity of the output voltage V_(O) 142 is coupled to thenon-inverting input of the voltage comparator 504 via a signal 530. Thesignal 530 receives the output voltage V_(O) 142 which is scaled by theresistors R3 506 and R4 508. In the depicted example, the voltagecomparator 504 outputs a logic high if the output voltage V_(O) 142 isgreater than or equal to the desired regulated voltage V_(REG). In oneexample, when output voltage V_(O) 142 is greater than or equal to thedesired regulated voltage V_(REG), a logic high output of the voltagecomparator 504 allows a single delayed enable pulse at the output 522 ofthe synchronous rectifier controller 502 to trigger the monostablemultivibrator 510. In other words, the synchronous rectifier controller502 outputs a delayed enable pulse 522 to activate the monostablemultivibrator 510 when the output of the voltage comparator 504 is atlogic high. The monostable multivibrator circuit 510 further generates asingle pulse which is inverted using the inverter 512. The synchronousrectifier controller 502 is designed to sense a secondary voltage V_(S)148 and I_(S) 138 greater than zero using a secondary current sensesignal 144.

When the primary switch S1 136 turns off, the secondary voltage V_(S)148 is positive with respect to the output return 130 and secondarycurrent I_(S) 138 is greater than zero. The secondary current I_(S) 138is in the direction from the secondary winding 110 through the switchedelement 122 to the load 128. The synchronous rectifier controller maysense when the primary switch S1 136 turns off by sensing a positivesecondary voltage V_(S) 148 and/or a positive secondary current I_(S)138. The synchronous rectifier controller 502 is coupled to turn ontransistor 306 in the switched element 122 when the primary switch S1136 turns off. The synchronous rectifier controller 502 outputs theenable signal 516 to turn on transistor 306 in the switched element 122.However, the output of the AND gate 514 has to be at a logic high valuewith respect to input 162 in order for transistor 306 to be turned on.The AND gate 514 outputs a logic high value when both signals 516 and518 are logic high. However, the signal 518 is momentarily logic lowwhen the output voltage V_(O) 142 is greater than the desired regulatedvoltage V_(REG). As such, the transistor 306 is turned off momentarilywhen the output voltage V_(O) 142 is greater than or equal to thedesired regulated voltage V_(REG). In other words, the turning on of thetransistor 306 is inhibited for a predetermined amount of time (theduration of the disable pulse on output 532) when the scaled outputvoltage V_(O) 142 at node 530 is greater than the desired regulatedvoltage V_(REG). As illustrated later in the depiction of a secondarydrive signal 123 in FIG. 6 and FIG. 7, the duration of the disable pulseon output 532 is shorter than the duration of the enable signal 516. Inone example, the duration of the disable pulse may provide informationabout output current, or temperature, or any other quantity which mayrepresent the load condition. In the example power converter 100 of FIG.1, the duration of the disable pulse may be proportional to the outputcurrent that is, the duration of the disable pulse may be longer forlower output current and shorter for higher output current. In otherwords, in the example power converter 100 of FIG. 1, the duration of thedisable pulse may be longer for lighter loads and shorter for heavierloads. In another embodiment the duration of the disable pulse may beproportional to the temperature of the power converter; that is; theduration of the disable pulse may be longer for lower temperatures andshorter for higher temperatures. In yet another embodiment, the durationof the disable pulse may be proportional to another quantity indicatingthe load condition. In yet another embodiment the combination of delaytime and the duration of disable pulse may provide information aboutoutput current, or temperature, or any other quantity which mayrepresent the load condition.

FIG. 5B is a functional block diagram 500B that illustrates details ofanother example secondary control circuit 124 of the power converter 100in accordance with the teachings of the present invention. The secondarycontrol circuit 124 of 500B is comprised of the same circuit elements asthe secondary control circuit 124 of 500A arranged such that elements ofdiagram 500A have the same voltages and conduct the same currents as theelements of diagram 500B. Therefore, the secondary control circuit 124of 500B works substantially in a similar manner as the secondary controlcircuit 124 of 500A. A difference between the secondary control circuit124 of 500A and 500B is that in 500B, the input 162 of the switchedelement 122 is coupled to the output return 130 and the output 164 ofthe switched element 122 is coupled to the secondary winding N_(S) 110.

FIG. 6 is a timing diagram 600 that shows waveforms illustratingoperation of the power converter 100 in discontinuous conduction mode(DCM) with the switched element 122 configured as illustrated by eitherFIG. 5A or FIG. 5B. In a typical DCM operation, the current in thesecondary winding may be substantially zero before the primary switchturns on. According to the configuration of the switched element 122 inthe depicted example, the synchronous rectifier has the characteristicsof FIG. 3 with diode 304 having one value of impedance and transistor306 having another value of impedance. As stated before, the switchedelement voltage V_(SR) 160 is V_(T1) and V_(D1) when the secondary drivesignal 123 is high and low respectively due to the change in theimpedances. The change in the switched element voltage V_(SR) 160 fromV_(T1) to V_(D1) is reflected at the bias winding voltage V_(B) 137.

Two complete switching periods T_(S1) 602 and T_(S2) 612 are illustratedfor various waveforms in the timing diagram 600 for the power converter100 in FIG. 1 operating in DCM. The switching period T_(S1) 602illustrates the operation of the power converter 100 with onlytransistor 306 substantially conducting the secondary current I_(S) 138in the secondary winding 110, whereas the switching period T_(S2) 612illustrates the operation of the power converter 100 with the transistor306 and the diode 304 conducting the secondary current I_(S) 138 in thesecondary winding 110 at different times. During the switching periodT_(S1) 602, the output voltage V_(O) 142 is less than the desiredregulated value V_(REG); therefore, the secondary drive signal 123remains high for the entire time when there is the secondary currentI_(S) 138 in the secondary winding 110. During the switching periodT_(S2) 612, the output voltage V_(O) 142 may be greater than or equal tothe desired regulated value V_(REG); therefore, the secondary drivesignal 123 becomes low for the interval T4 624 when there is thesecondary current I_(S) 138 in the secondary winding 110.

The primary drive signal 152 from primary control circuit 114 is highfor duration T_(ON1) 604. During T_(ON1) 604 the primary switch S1 136conducts primary switch current ID 134. During the interval T_(ON1) 604,the primary switch current ID 134 increases until the primary drivesignal 152 turns the primary switch S1 136 off at the end of theinterval T_(ON1) 604. During the interval T_(ON1) 604, the secondaryvoltage V_(S) 148, the bias winding voltage V_(B) 137, and the switchedelement voltage V_(SR) 160 are negative. The primary drive signal 152 islow for duration T_(OFF1) 606, preventing the primary switch S1 136 fromconducting current. Once the primary switch S1 136 is off, the switchedelement 122 conducts the secondary current I_(S) 138 in secondarywinding 110. During the interval T₁ 608, the secondary voltage V_(S) 148and the secondary current I_(S) 138 are positive. At the beginning of T₁608, the secondary control circuit 124 makes the secondary drive signal123 high. The secondary drive signal 123 from secondary control circuit124 remains high for the duration T₁ 608 with transistor 306substantially conducting the secondary current I_(S) 138. The switchedelement voltage V_(SR) 160 is substantially equal to the voltage V_(T1)across the transistor 306. During the interval T₁ 608, the currentthrough the diode 304 is substantially zero. It is appreciated that inpractice, there may be some leakage current through the diode 304 duringthe interval T₁ 608. During this time, the secondary voltage V_(S) 148is substantially equal to the sum of output voltage V_(O) 142 and thevoltage V_(T1) across the transistor 306. The bias winding voltage V_(B)137 is substantially proportional to the secondary voltage V_(S) 148during the interval T₁ 608. The secondary current I_(S) 138 graduallydecreases to zero at the end of T₁ 608. The rate of decrease of thesecondary current I_(S) 138 is substantially constant throughout theinterval T_(OFF1) 606. The secondary current I_(S) 138 remains zero fromthe time t_(X1) 610 until the beginning of the next switching periodT_(S2) 612. The secondary voltage V_(S) 148 and bias winding voltageV_(B) 137 decrease to zero at time t_(X1) 610 and remain zero until thebeginning of the next switching period T_(S2) 612.

The primary drive signal 152 becomes high again at the beginning ofinterval T_(S2) 612 for a duration T_(ON2) 614 allowing the primaryswitch S1 136 to conduct the primary switch current ID 134. During theinterval T_(ON2) 614, all the signals shown in the waveforms of FIG. 6behave similarly as discussed with respect to the duration T_(ON1) 604of FIG. 6. At the end of T_(ON2) 614, the primary drive signal 152becomes low preventing the primary switch S1 136 from conducting theprimary switch current I_(D) 134. At the beginning of T₂ 618, thesecondary current I_(S) 138 increases suddenly. During the interval T₂618, the secondary control circuit 124 makes the secondary drive signal123 high allowing the transistor 306 to conduct the secondary currentI_(S) 138. During this time, the secondary voltage V_(S) 148 and thebias winding voltage V_(B) 137 are positive. The secondary voltage V_(S)148 is substantially equal to the sum of the output voltage V_(O) 142and the voltage V_(T1) across the transistor 306. The bias windingvoltage V_(B) 137 is substantially proportional to the secondary voltageV_(S) 148.

During the switching period T_(S2) 612, the output voltage V_(O) 142 maybe greater than or equal to the desired regulated value V_(REG) which issensed by voltage comparator 504. The output of the voltage comparator504 then becomes high, thereby allowing delayed enable pulse 522 totrigger the monostable multivibrator 510. The secondary drive signal 123then becomes low in response to the output of the inverter 518 for theinterval T₄ 624, turning the transistor 306 off and preventing thetransistor 306 from conducting any secondary current I_(S) 148. Duringthe interval T₄ 624, the transistor 306 remains off and the diode 304substantially conducts the secondary current I_(S) 138. During this timethe switched element voltage V_(SR) 160 is substantially equal to theforward voltage V_(D1) across the diode 304. During this time, thecurrent through the transistor 306 is substantially zero. It isappreciated that in practice, there may be some leakage current throughthe transistor 306 during the interval T₄ 624. At the end of interval T₂618, there is a sudden increase in the secondary winding voltage V_(S)148. During the interval T₄ 624, the secondary winding 110 has asecondary voltage V_(S) 148 substantially equal to the sum of the outputvoltage V_(O) 142 and the voltage V_(D1) across the diode 304. The biaswinding voltage V_(B) 137 also increases during the interval T4 624 asthe voltage pulse is reflected onto bias winding 108 from secondarywinding 110. From the end of interval T₃ 620 until the time t_(X2) 622,the secondary drive signal 123 becomes high again allowing thetransistor 306 to substantially conduct the secondary current I_(S) 148with the diode 304 conducting substantially zero current; however, theremay be some leakage current through the diode 304 from the end ofinterval T₃ 620 until the time t_(X2) 622. From the end of interval T₃620 until the time t_(X2) 622, the secondary voltage V_(S) 148 decreasesslightly and is substantially equal to the sum of the output voltageV_(O) 142 and the voltage V_(T1) across the transistor 306.

At time t_(X2) 622 the secondary current I_(S) 138, the secondaryvoltage V_(S) 148, and the bias winding voltage V_(B) 137 decrease tozero. The secondary current I_(S) 138 suddenly increases to a non-zerovalue at the beginning of interval T₂ 618 and slowly decreases to zeroat the time t_(X2) 622. The rate at which the secondary current I_(S)138 decreases varies during the period T_(OFF2) 616 for high and lowvalues of the secondary drive signal 123. This is due to the fact thatswitched element voltage V_(SR) 160 varies during the period T_(OFF2)616 for high and low values of the secondary drive signal 123. Thesecondary current I_(S) 138 decreases at a first rate for the durationof T₂ 618 and from the end of interval T₃ 620 until t_(X2) 622 when thetransistor 306 is substantially conducting. The secondary current I_(S)138 decreases at a second rate that is greater than the first rateduring the interval T₄ 624, when the diode 304 is substantiallyconducting. This is because the forward voltage V_(D1) across the diode304 is greater than the forward voltage V_(T1) across the transistor 306which causes the switched element voltage V_(SR) 160 to vary during theinterval T_(OFF2) 616.

In response to the increase in the bias winding voltage V_(B) 137 duringthe interval T₄ 624, the primary switch S1 136 may be on or off for thenext switching period. In other words, at the end of the switchingperiod T_(S2) 612, the primary drive signal 152 may be high or low asdepicted by a dashed line at the end of the switching period T_(S2) 612in accordance with the control algorithm.

FIG. 7 is a timing diagram 700 that shows waveforms illustratingoperation of the power converter 100 when it operates in a continuousconduction mode (CCM) with the switched element 122 configured asillustrated by either FIG. 5A or FIG. 5B. In a typical CCM operation,there may be current in the secondary winding when the primary switchturns on. According to the configuration of the switched element 122 inthe depicted example, the synchronous rectifier has the characteristicsof FIG. 3 with diode 304 having one value of impedance and transistor306 having another value of impedance. As stated before, the switchedelement voltage V_(SR) 160 is V_(T1) and V_(D1) when the secondary drivesignal 123 is high and low respectively due to the change in theimpedances. The change in the switched element voltage V_(SR) 160 fromV_(T1) to V_(D1) is reflected at the bias winding voltage V_(B) 137. Onedifference between the DCM and CCM is that when the power converter 100in FIG. 1 operates in CCM, there is energy in energy transfer element T1104 during the entire time that primary switch S1 136 is open. In otherwords, there is energy in energy transfer element T1 104 when primaryswitch S1 136 initially closes. When the power converter 100 in FIG. 1operates in CCM, the primary switch current I_(D) 134 in primary switchS1 136 has a value greater than zero immediately after primary switch S1136 closes.

Two complete switching periods T_(S1) 702 and T_(S2) 712 are illustratedfor various waveforms in the timing diagram 700 for the power converter100 in FIG. 1 operating in CCM. The switching period T_(S1) 702illustrates the operation of the power converter 100 with only thetransistor 306 substantially conducting the secondary current I_(S) 138in the secondary winding 110, whereas the switching period T_(S2) 712illustrates the operation of the power converter 100 with the transistor306 and the diode 304 conducting the secondary current I_(S) 138 in thesecondary winding 110 at different times. During the switching periodT_(S1) 702, the output voltage V_(O) 142 is less than the desiredregulated value V_(REG), therefore, the secondary drive signal 123remains high for the entire time when there is the secondary currentI_(S) 138 in the secondary winding 110. During the switching periodT_(S2) 712, the output voltage V_(O) 142 may be greater than or equal tothe desired regulated value V_(REG), therefore, the secondary drivesignal 123 becomes low for the interval T₄ 724, when there is secondarycurrent I_(S) 138 in the secondary winding 110.

The primary drive signal 152 from primary control circuit 114 is highfor duration T_(ON1) 704, allowing the primary switch S1 136 to conductprimary switch current I_(D) 134. During the interval T_(ON1) 704, theprimary switch current I_(D) 134 increases until the primary drivesignal 152 turns the primary switch S1 136 off at the end of theinterval T_(ON1) 704. During the interval T_(ON1) 704, the secondaryvoltage V_(S) 148, the bias winding voltage V_(B) 137, and the switchedelement voltage V_(SR) 160 are negative. The primary drive signal 152 islow for duration T_(OFF1) 706, preventing the primary switch S1 136 fromconducting primary switch current ID 134 and allowing the switchedelement 122 to conduct the secondary current I_(S) 138 in secondarywinding 110. During the interval T₁ 708, the secondary voltage V_(S) 148and the secondary current I_(S) 138 are positive. At the beginning of T₁708, the secondary control circuit 124 makes the secondary drive signal123 high. The secondary drive signal 123 from secondary control circuit124 remains high for the duration T₁ 708 with transistor 306substantially conducting the secondary current I_(S) 138. The switchedelement voltage V_(SR) 160 is substantially equal to the voltage V_(T1)across the transistor 306. During the interval T₁ 708 the currentthrough the diode 304 is substantially zero. It is appreciated that inpractice, there may be some leakage current through the diode 304 duringthe interval T₁ 708. During this time, the secondary voltage V_(S) 148is substantially equal to the sum of output voltage V_(O) 142 and thevoltage V_(T1) across the transistor 306. The bias winding voltage V_(B)137 is substantially proportional to the secondary voltage V_(S) 148during the interval T₁ 708. The secondary current I_(S) 138 graduallydecreases during T₁ 708 but may not necessarily reach zero before thebeginning of the switching period T_(S2) 712. The rate of decrease ofthe secondary current I_(S) 138 is substantially constant throughout theinterval T_(OFF1) 706. The secondary voltage V_(S) 148 and bias windingvoltage V_(B) 137 may not necessarily decrease to zero before thebeginning of the next switching period T_(S2) 712.

The primary drive signal 152 becomes high again at the beginning ofinterval T_(S2) 712 for a duration T_(ON2) 714 allowing the primaryswitch S1 136 to conduct the primary switch current I_(D) 134. Duringthe interval T_(ON2) 714, all the signals shown in the waveforms of FIG.7 behave similarly as discussed with respect to the duration T_(ON1) 704of FIG. 7. At the end of T_(ON2) 714, the primary drive signal 152becomes low preventing the primary switch S1 136 from conducting theprimary switch current I_(D) 134. At the beginning of T₂ 718, thesecondary current I_(S) 138 increases suddenly. During the interval T₂718, the secondary control circuit 124 makes the secondary drive signal123 high allowing the transistor 306 to conduct the secondary currentI_(S) 138. During this time, the secondary voltage V_(S) 148 and thebias winding voltage V_(B) 137 are positive. The secondary voltage V_(S)148 is substantially equal to the sum of output voltage V_(O) 142 andthe voltage V_(T1) across the transistor 306. The bias winding voltageV_(B) 137 is substantially proportional to the secondary voltage V_(S)148.

During the switching period T_(S2) 712, the output voltage V_(O) 142 maybe greater than or equal to the desired regulated value V_(REG) which issensed by the voltage comparator 504. The output of the voltagecomparator 504 then becomes high, thereby allowing delayed enable pulse522 to trigger the monostable multivibrator 510. The secondary drivesignal 123 then becomes low in response to the output of the inverter518 for the interval T₄ 724, turning the transistor 306 off andpreventing the transistor 306 from conducting any secondary currentI_(S) 148. During the interval T₄ 724, the transistor 306 remains offand the diode 304 substantially conducts the secondary current I_(S)138. During the interval T₄ 724, the switched element voltage V_(SR) 160is substantially equal to the forward voltage V_(D1) across the diode304 and the current through the transistor 306 is substantially zero. Itis appreciated that in practice, there may be some leakage currentthrough the transistor 306 during the interval T₄ 724. At the end ofinterval T₂ 718, there is a sudden increase in the secondary windingvoltage V_(S) 148. During the interval T₄ 724, the secondary voltageV_(S) 148 is substantially equal to the sum of the output voltage V_(O)142 and the voltage V_(D1) across the diode 304. The bias windingvoltage V_(B) 137 also increases during the interval T₄ 724 in responseto the sudden increase in the secondary voltage V_(S) 148. From the endof interval T3 720 until the end of interval T_(OFF2) 716, the secondarydrive signal 123 becomes high again allowing the transistor 306 tosubstantially conduct the secondary current I_(S) 148 with the diode 304conducting substantially zero current, however, it is appreciated thatin practice, there may be some leakage current through the diode 304during this time. From the end of interval T₃ 720 until the end ofinterval T_(OFF2) 716, the secondary voltage V_(S) 148, decreasesslightly and is substantially equal to the sum of the output voltageV_(O) 142 and the voltage V_(T1) across the transistor 306. At the endof interval T_(OFF2) 716 the secondary current I_(S) 138, the secondaryvoltage V_(S) 148, and the bias winding voltage V_(B) 137 may be stillpositive. The secondary current I_(S) 138 suddenly increases to anon-zero value at the beginning of interval T₂ 718 and decreases untilthe end of interval T_(OFF2) 716. The secondary current I_(S) 138 maynot necessarily decrease to zero before the beginning of the nextswitching period (immediately following interval T_(OFF2) 716).

The rate at which the secondary current I_(S) 138 decreases variesduring the period T_(OFF2) 716 for high and low values of the secondarydrive signal 123. This is due to the fact that switched element voltageV_(SR) 160 varies during the period T_(OFF2) 716 for high and low valuesof the secondary drive signal 123. The secondary current I_(S) 138decreases at a first rate for the duration of T₂ 718 and from the end ofinterval T₃ 720 until the end of interval T_(OFF2) 716 when thetransistor 306 is substantially conducting. The secondary current I_(S)138 decreases at a second rate that is greater than the first rateduring the interval T₄ 724, when the diode 304 is substantiallyconducting. This is because the forward voltage V_(D1) across the diode304 is greater than the forward voltage V_(T1) across the transistor 306which causes the switched element voltage to vary during the intervalT_(OFF2) 716.

In response to the increase in the bias winding voltage V_(B) 137 duringthe interval T₄ 724, the primary switch S1 136 may be on or off for thenext switching period. In other words, at the end of the switchingperiod T_(S2) 712, the primary drive signal 152 may be high or low asdepicted by a dashed line at the end of the switching period T_(S2) 712in accordance with the control algorithm.

It is appreciated, however, when operating in CCM, the current I_(S) 138in the secondary winding 110 may not necessarily reduce to zero in allexamples when a switching cycle is skipped. It may take several skippedswitching cycles before the current I_(S) 138 in the secondary winding110 reduces to zero. The example waveforms of FIG. 7 exaggerate theslope of the current I_(S) 138 in the secondary winding 110 forillustration, but in practice the slope is likely to be much lessrelative to the peak current when operating in CCM.

It should be noted that in the example power converter 100 of FIG. 1some switching cycles may be CCM and others may be DCM.

FIG. 8 is a flow diagram 800 that illustrates an example process forcontrolling the primary switch S1 136 by inducing a voltage change atthe secondary winding 110 and observing the change in the voltage at thebias winding 108 so that the primary control circuit 114 may observewhether the output voltage V_(O) 142 is within a desired range. Theorder in which some or all of the process blocks appear in flow diagram800 should not be deemed limiting. Rather, one of ordinary skill in theart having the benefit of the present disclosure will understand thatsome of the process blocks may be executed in a variety of orders notillustrated, or even in parallel.

After starting at block 801, at block 802 the primary switch S1 136 canbe turned on and then off. At the end of block 802 the process proceedsto block 803.

At block 803, it can be checked if the output voltage V_(O) 142 isgreater than or equal to the desired regulated voltage V_(REG). If theoutput voltage V_(O) 142 is greater than or equal to the desiredregulated voltage V_(REG), then the process proceeds to block 804. Ifoutput voltage V_(O) 142 is less than the desired regulated voltageV_(REG), then process 803 will proceed to block 808. At block 808, theprocess may wait until the beginning of the next switching period. Whenthe next switching period occurs, the process will go back to thebeginning of block 802.

At block 804, it can be checked if there is current in the secondarywinding 110. If there is current I_(S) 138 in the secondary winding 110,then the process proceeds to block 805. If there is no current in thesecondary winding 110 then the process proceeds to block 808. At block808, the process may wait until the beginning of the next switchingperiod. When the next switching period occurs, the process will go backto the beginning of block 802.

At block 805, the secondary voltage V_(S) 148 can be changed by turningoff the transistor 306 momentarily to alter the voltage at the secondarywinding 110 to induce a voltage change at the bias winding 108 so thatthe primary control circuit knows whether the output voltage V_(O) 142is greater than or equal to the desired regulated value V_(REG). At theend of block 805, the process proceeds to block 806.

At block 806, the voltage change at the secondary winding 110 can beobserved at the bias winding 108 by the primary control circuit 114 toknow whether the output voltage V_(O) 142 is greater than or equal tothe desired regulated value V_(REG). In the example power converter 100of FIG. 1, the voltage change at the bias winding 108 can be observedusing the voltage observer circuit 402. At the end of block 806, theprocess proceeds to block 807.

At block 807, the switching of the primary switch S1 136 can be delayedfor one or more switching periods based on the control algorithm. In theexample power converter 100 of FIG. 1, the primary switch S1 136 can beturned off by controlling the primary drive signal 152.

At the end of block 807, the process goes back to the beginning of block802 to check if the output voltage V_(O) 142 is greater than or equal tothe desired regulated voltage V_(REG).

FIG. 9 is a flow diagram 900 that illustrates an example process forregulating a power converter in accordance with the teachings of thepresent invention. The order in which some or all of the process blocksappear in flow diagram 900 should not be deemed limiting. Rather, one ofordinary skill in the art having the benefit of the present disclosurewill understand that some of the process blocks may be executed in avariety of orders not illustrated, or even in parallel.

At block 901, a switched element (e.g. switched element 122) is enabledfor an enabling period (e.g. 602, 702, 612, or 712) in response tosensing that primary switch S1 136 is off. Sensing that primary switchS1 136 is off may be accomplished by sensing the secondary current I_(S)138 through secondary winding 110, by sensing the secondary voltageV_(S) 148 across secondary winding 110, or some combination of the two.At block 902, a voltage pulse on the secondary winding 110 is generatedduring a disabling period (e.g. voltage increase on V_(S) 148 during T₄624/724) in response to the output V_(O) 142 of the power converterbeing at or above a reference value. The voltage pulse on the secondarywinding 110 is generated by adjusting the voltage V_(SR) 160 acrossswitched element 122 while it conducts current I_(O) 140 to an output.At block 903, primary switch S1 136 is switched in response to thevoltage pulse being reflected onto bias winding 108. Primary controller114 may skip a switching cycle of primary switch S1 136 in response tosensing the voltage pulse on bias winding 108. The flow may return toprocess block 901 at the conclusion of process block 903.

The above description of illustrated embodiments of the invention,including what is described in the Abstract, is not intended to beexhaustive or to limit the invention to the precise forms disclosed.While specific embodiments of, and examples for, the invention aredescribed herein for illustrative purposes, various modifications arepossible within the scope of the invention, as those skilled in therelevant art will recognize.

These modifications can be made to the invention in light of the abovedetailed description. The terms used in the following claims should notbe construed to limit the invention to the specific embodimentsdisclosed in the specification. Rather, the scope of the invention is tobe determined entirely by the following claims, which are to beconstrued in accordance with established doctrines of claiminterpretation.

What is claimed is:
 1. A power converter comprising: An energy transferelement including a first winding, a second winding, and a thirdwinding, wherein the first winding is coupled to an input voltage andthe second winding is coupled to an output of the power converter; aswitched element coupled to the second winding; a secondary controlcircuit coupled to generate a voltage pulse across the second windingwhile the second winding provides current to the output, wherein thesecondary control circuit is coupled to vary a voltage across theswitched element to generate the voltage pulse across the second windingin response to the output of the power converter being at or above areference value; a primary switch coupled to the first winding; and aprimary control circuit coupled to the primary switch and the thirdwinding, the primary control circuit coupled to switch the primaryswitch to regulate the output of the power converter in response to thevoltage pulse, wherein the second winding is coupled to reflect thevoltage pulse onto the third winding.
 2. The power converter of claim 1,wherein the switched element includes a transistor and the secondarycontrol circuit is coupled to output a secondary drive signal toselectively disable the transistor to increase the voltage across theswitched element while a secondary current through the second windingprovides current to the output in response to the output of the powerconverter falling below the reference value.
 3. The power converter ofclaim 2, wherein the switched element also includes a diode coupled inparallel with the transistor, wherein the diode has a forward voltagethat is larger than a transistor voltage across the transistor when thetransistor is enabled.
 4. The power converter of claim 2, wherein thesecondary control circuit includes: a first comparator coupled togenerate an output signal on a comparator output when the output of thepower converter is above the reference value; a synchronous rectifiercontroller coupled to sense at least one of a secondary voltage acrossthe second winding and a secondary current through the second winding todetermine when the primary switch is off, wherein the synchronousrectifier controller is configured to generate an enable signal and adelayed enable pulse when the secondary voltage and/or the secondarycurrent indicates that the primary switch is off; a pulse generatorcoupled to generate a disable pulse at a pulse generator output inresponse to receiving the output signal and the delayed enable pulse,wherein the disable pulse has a shorter duration than a duration of theenable signal; and secondary logic coupled to generate the secondarydrive signal in response to the enable signal and the disable pulse,wherein the secondary drive signal disables the transistor when thesecondary logic receives the enable signal and the disable pulse, andwherein the secondary drive signal enables the transistor when thesecondary logic receives the enable signal and not the disable pulse. 5.The power converter of claim 4, wherein the pulse generator includes: anAND gate coupled to receive the delayed enable pulse and the outputsignal; and a monostable multivibrator coupled to generate the disablepulse in response to an output of the AND gate.
 6. The power converterof claim 4, wherein the delayed enable pulse is generated after a delaytime, and wherein the delay time indicates at least one of an outputcurrent or a temperature of the power converter.
 7. The power converterof claim 4, wherein the delayed enable pulse is generated after a delaytime, and wherein the delay time increases when an output current of thepower converter changes in a first direction, and wherein the delay timedecreases when the output current changes in a second direction that isopposite the first direction.
 8. The power converter of claim 4, whereina duration of the delayed enable pulse indicates at least one of anoutput current or a temperature of the power converter.
 9. The powerconverter of claim 4, wherein a duration of the delayed enable pulseincreases when an output current of the power converter changes in afirst direction, and wherein the duration of the delayed enable pulsedecreases when the output current changes in a second direction that isopposite the first direction.
 10. The power converter of claim 2,wherein a magnitude of rate of change of the secondary current throughthe second winding increases when the voltage pulse is across the secondwinding.
 11. The power converter of claim 1, wherein the primary controlcircuit is coupled to delay switching the primary switch for one or moreperiods in response to observing the voltage pulse.
 12. The powerconverter of claim 1, wherein the primary control circuit includesvoltage sense circuitry coupled to the third winding to observe thevoltage pulse.
 13. The power converter of claim 1, wherein the secondwinding of the energy transfer element is galvanically isolated from thefirst and third windings.
 14. The power converter of claim 1, whereinthe power converter is a flyback converter topology.
 15. The powerconverter of claim 1, wherein the energy transfer element includes atransformer.
 16. The power converter of claim 1, wherein the energytransfer element includes a coupled inductor.
 17. A controller for apower converter comprising: a synchronous rectifier controller coupledto sense a secondary current through a secondary winding of the powerconverter, wherein the synchronous rectifier controller is configured togenerate an enable signal and a delayed enable pulse when the secondarycurrent increases from substantially zero to a threshold current value;a comparator coupled to generate an output signal on a comparator outputwhen an output of the power converter is above a reference value; apulse generator coupled to generate a disable pulse at a pulse generatoroutput in response to receiving the output signal and the delayed enablepulse, wherein the disable pulse has a shorter duration than a durationof the enable signal; and secondary logic to be coupled to a switchedelement coupled to the secondary winding, wherein the secondary logic isconfigured to generate a secondary drive signal to vary a voltage acrossthe switched element to create a voltage pulse across the secondarywinding.
 18. The controller of claim 17, wherein the switched elementincludes a transistor and the secondary logic is coupled to output thesecondary drive signal to selectively disable the transistor to increasethe voltage across the switched element while the secondary windingprovides current to the output in response to the output of the powerconverter falling below the reference value.
 19. The controller of claim18, wherein the secondary drive signal disables the transistor when thesecondary logic receives the enable signal and the disable pulse, andwherein the secondary drive signal enables the transistor when thesecondary logic receives the enable signal and not the disable pulse.20. The controller of claim 17, wherein the pulse generator includes: anAND gate coupled to receive the delayed enable pulse and the outputsignal; and a monostable multivibrator coupled to generate the disablepulse in response to an output of the AND gate.
 21. The controller ofclaim 17, wherein the voltage pulse across the secondary windinginfluences switching of a primary switch coupled to a primary windinginductively coupled to the secondary winding, the secondary drive signalgenerated in response to receiving the enable signal and the disablepulse.
 22. A method of regulating a power converter, comprising:enabling a switched element for an enabling window in response tosensing that a primary switch of the power converter is off, wherein theswitched element is coupled to a second winding of a an energy transferelement that also includes a third winding and a first winding coupledto an input of the power converter, and wherein the primary switch iscoupled to the first winding to regulate an output of the powerconverter; generating a voltage pulse on the second winding by adjustinga voltage across the switched element for a disabling duration inresponse to the output of the power converter being at or above areference value, wherein the disabling duration is within the enablingwindow, and wherein secondary current is provided to the output throughthe second winding during the enabling window; and switching the primaryswitch in response to the voltage pulse being reflected onto the thirdwinding, wherein the second winding is coupled to reflect the voltagepulse onto the third winding.
 23. The method of claim 22, wherein theswitched element includes a transistor and a diode, and wherein enablingthe switched element includes enabling the transistor.
 24. The method ofclaim 23, wherein adjusting the voltage across the switched element forthe disabling duration includes disabling the transistor for thedisabling duration.
 25. The method of claim 23, wherein the secondwinding of the energy transfer element is galvanically isolated from thefirst and third windings.
 26. The method of claim 23, wherein saidsensing that the primary switch of the power converter is off includessensing secondary current through the second winding.
 27. The method ofclaim 26, further comprising: waiting until a next switching period ofthe primary switch in response to no secondary current being sensed inthe second winding; and switching the primary switch on and off at thenext switching period.
 28. The method of claim 23, wherein said sensingthat the primary switch of the power converter is off includes sensing asecondary voltage across the second winding.